This invention relates generally to radar processing systems, and more particularly to a pulse compression system used therein.
In the art of pulse radar systems, it is well known that the ability of a radar system to perform detection depends upon the energy content of transmitted pulses. The larger the energy content, the higher the signal-to-noise ratio of the returning echoes. A large energy content may be obtained through pulses with large peak power and/or long pulse duration. Pulses with small durations or pulse widths are preferred since the shorter the pulse width the better the range resolution and range accuracy. The components comprising a radar system, however, often place limits on the peak power of a pulse and require that pulses of longer duration be transmitted in order to obtain the necessary energy content for a pulse. Pulse compression techniques and pulse compressors were developed to achieve the same resolution of a much shorter pulse with high peak power using much longer pulses with lower peak power.
The use of pulse compression techniques has become increasingly more important because pulse compression has made the use of solid state devices in radar applications feasible. While not being able to produce short pulses with large peak power, solid state devices can produce lower powered pulses of long duration.
Initially pulse compressors were implemented as analog devices. These devices, however, were inflexible with respect to pulse length. To change the pulse length required changing the device. Additionally, the components of analog pulse compressors experienced gain drift due to changes in temperature and aging. Furthermore, the manufacture of some of these devices was complicated by the fact that each device produced differed from the other due to deviations during the manufacturing process. This required additional steps to align each device.
The disadvantages above drove the production of digitally implemented pulse compressors. Digitally implemented pulse compressors eliminated many of the disadvantages of the analog pulse compressors such as inflexible pulse lengths. While initially these systems were very hardware intensive and expensive to produce, at the present time they are relatively inexpensive and easy to implement.
An important consideration in the design of digital systems is the sample rate of the signal being processed. This rate must be high enough to ensure that the signal is represented accurately without distortion due to aliasing. At the same time, the higher the rate, the greater the processing throughput required of the digital hardware, and hence the greater the cost. Thus it is desirable to minimize the sampling rate under the constraint of not introducing distortion which degrades the performance of the system.
Few digital pulse compression system designs constrain or achieve a constrained sampling rate. U.S. Pat. No. 4,404,562 to Kretschmer describes one such system. The Kretschmer system transmits a linear FM signal such as shown in FIG. 1 which has a frequency sweep (bandwidth) of F.sub.2 -F.sub.1 and performs pulse compression on the echoes by means of a digital pulse compressor 60 illustrated in FIG. 2.
In the Kretschmer device, a signal generator 10 generates an intermediate frequency signal having a frequency which varies linearly from a frequency F.sub.1 to a frequency F.sub.2 as time varies from an arbitrary time t.sub.0 to time t.sub.0 +T. This linearly frequency modulated waveform is supplied to the mixer 12 via line 14 wherein it is mixed up to radio frequency (RF) by heterodyning it with an RF signal supplied by an RF signal generator 16 via line 18. The resultant RF signal is amplified by a power amplifier 20, passed through a standard duplexer 22, and radiated from a radar antenna 24.
Echoes received by the antenna 24 are supplied by the duplexer 22 to a mixer 28 via a line 26. The mixer 28 beats or heterodynes the echo signal with the RF signals supplied by RF signal generator 16 via line 30 in order to obtain the intermediate frequency echo signal varying from F.sub.1 to F.sub.2. The resultant intermediate frequency signal is amplified in an intermediate frequency amplifier 32 with a bandwidth from F.sub.1 to F.sub.2 centered on the frequency (F.sub.2 +F.sub.1)/2. Since the intermediate frequency amplifier 32 passes the same band as the linear FM waveform, the intermediate frequency amplifier 32 performs noise reduction but does not perform a pulse compression function.
At this point in the circuit both the transmission and the reception processing have been analog in nature. The circuit then samples the echo signal at the Nyquist rate. The Nyquist rate is defined as twice the reciprocal of the frequency sweep or bandwidth of the linear FM waveform, or in this case 2/(F.sub.2 -F.sub.1). Where conversion to baseband I and Q signals is used, Kretschmer contends the Nyquist rate would be 1/(F.sub.2 -F.sub.1) for each baseband I and Q signal. However, an amplifier, such as intermediate frequency amplifier 32, providing no attenuation between F1 and F2 and high attenuation outside this band is not physically realizable. Consequently, sampling at 2/(F.sub.2 -F.sub.1) or 1/(F.sub.2 -F.sub.1) for each baseband I and Q signal is not adequate.
To obtain the information baseband, the intermediate frequency echo signal from intermediate frequency amplifier 32 is beat or heterodyned with a local oscillator (L.O.) intermediate frequency signal. Accordingly, an I channel and a Q channel are provided for generating baseband signals and sampling those signals at the Nyquist rate. The I channel comprises a multiplier 34 for beating or heterodyning the intermediate frequency echo signal on line 33 from amplifier 32 with an L.O. intermediate frequency signal from signal generator 10 via line 35. Likewise, a multiplier 36 in the Q channel multiplies the intermediate frequency echo signal on line 35 with an L.O. intermediate frequency signal from the signal generator 10 shifted in phase by 90.degree. by the phase shifter 38 and provided via line 40.
The baseband I and Q signals are then passed through low pass filters 42 and 44, respectively. Kretschmer discusses that these low pass filters may be optimally adjusted to just pass baseband pulses of length 1/(F.sub.2 -F.sub.1) (see col. 4 lines 30-35 of U.S. Pat. No. 4,404,562). In other words Kretschmer teaches setting the passbands of low pass filters 42 and 44 equal to the frequency sweep or bandwidth of the linear FM waveform. Accordingly since the low pass filters 42 and 44 pass the same band as the linear FM waveform, the low pass filters 42 and 44 perform noise reduction but do not perform a pulse compression function. The outputs of these low pass filters 42 and 44 are then applied to sample and hold circuits 46 and 48, respectively.
Kretschmer further teaches that to determine the optimum sampling rate for the sample and hold circuits 46 and 48 there are two competing factors which require consideration. In the ideal situation, sampling would begin at the beginning of the echo pulse. However, no provision in the circuit can be made for ensuring that sampling will begin at the beginning of the echo pulse. Accordingly, a sampling error of as much as 1/2 of a sampling period may exist. Kretschmer teaches reducing the sampling period or increasing the sampling rate so that the sampling error will be proportionately reduced (col. 4, lines 37-48 of U.S. Pat. No. 4,404,562). In the Kretschmer device, however, very high sampling rates reveal the sidelobe within 13 dB of the mainlobe, which is present in the typical linear FM response.
The Nyquist sampling rate is the minimum sampling rate which will allow the Kretschmer circuit to reconstruct all of the information for a given bandwidth. This rate is generally two times the reciprocal of the bandwidth, or in the case of baseband I and Q signals it is equal to the reciprocal of the bandwidth itself. Kretschmer teaches that the use of Nyquist rate sampling will provide an acceptable sampling error rate and will also provide a maximized mainlobe to sidelobe ratio (col. 4, lines 57-59 of U.S. Pat. No. 4,404,562). FIG. 5 illustrates the response of the Kretschmer system with over sampling. As FIG. 5 illustrates, the sidelobe is only 13 dB down from the mainlobe. When the Nyquist rate is used, as illustrated in FIG. 3, the sidelobe is 27 dB down from the mainlobe. As discussed above, the Nyquist rate is 1/(F.sub.2 -F.sub.1) for the baseband I and Q signals. Accordingly, the sample and hold circuits 46 and 48 are driven at this rate. The sampling pulses are supplied via line 50 from the signal generator 10. The sample and hold circuits 46 and 48 hold their sample values for a time 1/(F.sub.2 -F.sub.1) between the samples.
The I and Q samples from the sample and hold circuits 46 and 48 are used to modulate multipliers 52 and 54, respectively. An L.O. intermediate frequency signal is supplied to multiplier 52 in the I channel which will operate to modulate that intermediate frequency signal with the sampled I signal from the sample and hold circuit 46. Kretschmer teaches that this intermediate frequency signal may be the signal F.sub.1 supplied via line 35. Likewise, the multiplier 54 in the Q channel is supplied with an L.O. intermediate frequency signal in quadrature with the intermediate frequency signal supplied to the multiplier 52. This quadrature L.O. intermediate frequency signal is modulated by the sampled Q output signal from the sample and hold circuit 48. Again, Kretschmer teaches this L.O. intermediate frequency signal may be the L.O. intermediate frequency signal F.sub.1 shifted in phase by 90.degree. and supplied via line 40. The intermediate frequency signal output from the multipliers 52 and 54 are then added together by an addition circuit 56 and the sum supplied to a compression circuit 60. The purpose of the compression circuit 60 is to take successive samples in time, and weight those samples such that when a received signal is properly indexed in the circuit, its output will be a short pulse with a significant amplitude, i.e., a compression operation.
The compression circuit 60 is a tap delay line, whose length will, of course, be determined by the uncompressed length T of the transmitted pulse. The number of taps on the delay line is generally determined by the number of samples taken in the sample and hold circuits 46 and 48. The delay line is composed of a series of p-1 cascaded delay elements 62
each equal to a delay of .tau.=T/p wherein T/p =1(F.sub.2 -F.sub.1). A signal tap 64 is taken before each delay element 62 and a final tap 65 is taken after the last delay element for a total of p signal taps. The delay elements 62 may be formed by cable or standard RC transmission line cut to the proper length. Equal amounts of signal will be obtained from each tap by setting the tap impedances in the well known manner.
If the originally transmitted signal had contained a single frequency across the length of the pulse, then the signals from these p taps could be added without further processing. However, because the frequency varies with time during the length of the pulse, the signals on the individual signal taps must be progressively phase shifted back into phase. Accordingly, in order to bring the signals on the various signal taps into phase with each other, phase weighting elements 66, 68, 70 and 72 are provided for the different signal taps. The phase weights to be set in these individual phase weighting elements is determined in the well known manner as follows. For purposes of the present discussion, the phase of the signal on the last signal tap 65 will be taken as the reference. Accordingly, the phase weighting element 66 will provide a phase shift of zero. The phase shifts for the other phase weighting elements may be calculated as follows:
The frequency difference between the signals on any two taps f.sub.diff is ##EQU1##
For a linear frequency modulation, as in this instance, df/dt=a constant k or EQU f.sub.diff =kt (2)
also ##EQU2## .PHI. diff taps ##EQU3## The constant k is found to equal the slope of the frequency vs time transmission characteristic or (F.sub.2 -F.sub.1)/T=B/T. Thus, the phase weighting elements will provide the following phase shifts ##EQU4##
These phase weighting elements may comprise BNC cable or twisted pairs cut to the proper length in order to obtain the prescribed phase shifts.
When the linear variation of the frequency of the signal with time has been taken into account, then the output from the phase weighting elements 66 through 72 should all be in phase when a received echo pulse is properly indexed in the delay line. Accordingly, the weighted signal output from the phase weighting elements 66 through 72 are added together in an adding circuit 74.
As the discussion above reveals, the Kretschmer device may only be realized in the ideal world of computer simulations because the filter characteristics specified by Kretschmer are not physically realizable.
Furthermore, Kretschmer discusses that a sampling error of as much as 1/2 the sampling period may exist. FIG. 3 illustrates the response of the Kretschmer device when there is no sampling error, and FIG. 4 illustrates the response of the Kretschmer device when a sampling error of 1/2 a sampling period exists. A comparison of FIGS. 3 and 4 demonstrates that the peak response of the Kretschmer device degrades by 3.9 dB and the width of the main lobe widens significantly when such a sampling error exists. Therefore, as the sampling error increases, the Kretschmer device will be unable to resolve the closely spaced targets that were resolved when no sampling error was present. In the radar art, loss due to sampling error is called range sampling loss, and is often expressed as the difference between the peak signal-to-noise ratio and the signal-to-noise ratio averaged over all possible sampling points. Kretschmer's approach results in a range sampling loss of 1.5 dB.
As further discussed above, Kretschmer teaches that the performance of his device depends on the sampling rate. Specifically, reducing the sampling rate proportionally reduces the sampling error discussed above. FIG. 5 illustrates the deleterious affects of over sampling (sampling above the Nyquist rate) with the Kretschmer device. As illustrated in FIG. 5, the sidelobe is only 13 dB down from the mainlobe. When the Nyquist rate is used, as illustrated in FIG. 3, the sidelobe is 27 dB down from the mainlobe. The closer the sidelobe becomes to the mainlobe, the greater the chance of erroneous or missed target detection. For example, the sidelobes produced by clutter such as mountains or other targets could mask the mainlobe of a target and prevent detection of that target. Accordingly, Kretschmer as discussed previously mandates the use of the Nyquist sampling rate, the lowest possible sampling rate, as the sampling rate of the sample and hold circuits 46 and 48 so that the sidelobe weighting shown in FIG. 3 is obtained.
The Kretschmer device suffers from other disadvantages with respect to sidelobe suppression and the flexibility of the transmitted waveform. FIG. 6 illustrates the response of the Kretschmer device with sidelobe weighting, and is a copy of FIG. 7 in U.S. Pat. No. 4,404,562 discussed in col. 7 lines 36-42. Based on FIG. 6, the peak response of the Kretschmer device is about -3 dB, which represents a mismatch loss of 3 dB. As discussed previously, resolution and the range at which targets are detectable depend upon the energy content of the transmitted pulse. Consequently, the greater the mismatch loss, the greater the energy content of the transmitted pulse must be to overcome such a loss and obtain the desired range and resolution. A mismatch loss of 3 dB would require doubling the output power of the transmitter. In the real world, such power requirements are not economically feasible; especially with solid state devices. As such, a mismatch loss of 3 dB represents an unacceptably high mismatch loss for a practical system.
Additionally, the Kretschmer device can use only a linear FM waveform such as shown in FIG. 2 as the transmitted pulse. Consequently, the Kretschmer pulse compressor is not applicable or flexible enough to be used in a radar system using a non-linear FM waveform as the transmitted pulse.
Furthermore, Kretschmer claims that the device of U.S. Pat. No. 4,404,562 is Doppler tolerant. FIG. 7 illustrates the response of the Kretschmer device with Doppler. The Doppler chosen represents a high-speed target at L-band. A comparison of FIG. 7 to FIG. 3 shows a peak response reduction of 0.5 dB, and a significant broadening of the mainlobe. This, particularly the broadening of the mainlobe, adversely impacts the resolution of the Kretschmer system.